Torque Controller for Permanent Magnet Synchronous Motor

ABSTRACT

When output voltage V 1  of an electric power converter reaches a prescribed voltage V 1 * ref , a difference between V 1  and V 1 * ref  is integrated to correct a commanded torque to τo* (τ*=τo*+Δτ).

CLAIM OF PRIORITY

The present application claims priority from Japanese patent application serial No. 2009-058840, filed on Mar. 12, 2009, the content of which is hereby incorporated by reference into this application.

FIELD OF THE INVENTION

The present invention relates to a vector control system for a permanent magnet synchronous motor and, more particularly, to a technology for achieving highly stable torque control at quick response even near a point at which the output voltage of an electric power converter is limited (saturated).

BACKGROUND OF THE INVENTION

The output voltage of an electric power converter in a controller for controlling current supplied to an AC motor by vector control may be limited (saturated). Prior art applicable to this case is described in, for example, Japanese Patent Laid-open No. 2004-180441. Specifically, to prevent the AC motor from generating an overcurrent, a current command for the q-axis in the d-q coordinate system, which is an orthogonal rotational coordinate system, is corrected so that current Id in the d-axis direction becomes 0 and an interfering item of the q-axis, which interferes with the d-axis, is operated.

SUMMARY OF THE INVENTION

With ordinary industrial motors and AC servo motors, a ratio of a voltage drop due to a motor resistance to a DC voltage (several hundreds of volts) supplied to the electric power converter is small. In most cases, therefore, the output voltage of the electric power converter is not limited (saturated) in a low-speed range.

With motors mounted on vehicles, however, the motor resistance and the resistance of the harness that interconnects the electric power converter and motor are relatively large, so a ratio of a voltage drop due to the motor resistance and harness resistance to the DC voltage (several tens of volts) may become large. In this case, the output voltage of the electric power converter may be limited (saturated) even in the low-speed range.

Accordingly, adequate precision cannot be obtained from the calculation in equation (5) described on page 7 in Japanese Patent Laid-open No. 2004-180441 because the resistance R is omitted.

Even if a resistance is included in the equation, calculation precision is lowered in the low-speed range because the resistance changes by 20% to 30% due to the ambient temperature.

An object of the present invention is to provide a torque controller, for a permanent magnet synchronous motor, that is robust against motor constants and can achieve highly precise torque control at quick response.

Another object of the present invention is to provide a torque controller, for a permanent magnet synchronous motor, that can be used in a range in which a DC voltage supplied to the electric power converter is several volts to several hundreds of volts, that is robust against motor constants, and that can achieve highly precise torque control at quick response.

In an aspect of the present invention, when the output voltage of an electric power converter reaches a prescribed voltage, a commanded torque is corrected.

In a preferred embodiment of the present invention, when the output voltage of an electric power converter reaches a prescribed voltage, a difference between the prescribed voltage and the output voltage of the electric power converter is integrated and the integrated value is used to correct a commanded torque.

The preferred embodiment of the present invention can provide a vector controller, for a permanent magnet synchronous motor, that can achieve highly precise torque control at quick response even near a point at which the output voltage of an electric power converter is limited (saturated).

The preferred embodiment of the present invention can also provide a torque controller, for a permanent magnet synchronous motor, that can be applied to both an inexpensive current detecting system and a system from which a position sensor is eliminated.

Other objects and features of the present invention will be clarified in the embodiments described below.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor in a first embodiment of the present invention.

FIG. 2 specifically illustrates a vector control calculating part 9 a in the torque controller in FIG. 1.

FIG. 3 is a block diagram of a commanded torque correcting part 7 in the torque controller in FIG. 1.

FIG. 4 illustrates control characteristics under no load when a calculation for torque command correction is performed according to the first embodiment.

FIG. 5 illustrates control characteristics under load when a calculation for torque command correction is performed according to the first embodiment.

FIG. 6 illustrates another vector control calculation part 9 b.

FIG. 7 illustrates still another vector control calculation part 9 c.

FIG. 8 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor in a second embodiment of the present invention.

FIG. 9 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor in a third embodiment of the present invention.

FIG. 10 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor when the present invention is applied to an electric hydraulic pump system.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Embodiments of the present invention will be described below in detail with reference to the drawings.

First Embodiment

FIG. 1 is an exemplary block diagram showing the structure of a torque controller for a permanent magnet synchronous motor in a first embodiment of the present invention.

The main circuit in FIG. 1 includes a permanent magnet synchronous motor 1, an electric power converter 2 that outputs voltages in proportion to commanded three-phase voltages Vu*, Vv*, and Vw*, and a DC power supply 21 that supplies a DC voltage. Sensors included in the main circuit are a current detector 3 that detects three-phase AC currents iu, iv, and iw and a position sensor 4 that detects a position θ of the permanent magnet synchronous motor 1 by using a resolver or an encoder.

In the functional section, which is a control circuit, a frequency calculating part 5 generates a calculated frequency ω₁ from a detected positional value θc; a coordinate converting part 6 outputs detected currents Idc and Iqc for the d-axis and q-axis from detected values iuc, ivc, and iwc of the three-phase AC currents iu, iv, and iw and the detected positional value θc of the permanent magnet synchronous motor 1; a commanded torque correcting part 7 calculates a corrected commanded torque Δτ* by using a commanded torque τo* given from a high-end device, a difference between a prescribed voltage V₁*_(ref) and output voltage V₁ from the electric power converter 2, and a calculated frequency ω₁; a current command converting part 8 calculates commanded currents Id* and Iq* for the d-axis and q-axis by using the commanded torque τo* given from the high-end device and the corrected commanded torque Δτ*; a vector control calculating part 9 a calculates commanded voltages Vdc* and Vqc* for the d-axis and q-axis by using the command currents Id* and Iq* for the d-axis and q-axis, the detected currents Idc and Iqc, and the calculated frequency ω₁, in view of the electric constants of the permanent magnet synchronous motor 1; an output voltage calculating part 10 calculates the output voltage V₁ from the electric power converter 2 by using the commanded voltages Vdc* and Vqc* for the d-axis and q-axis; a coordinate converting part 11 calculates the commanded three-phase AC voltages Vu*, Vv*, and Vw*, which the electric power converter 2 should output, according to the commanded voltages Vdc* and Vqc* for the d-axis and q-axis and the detected positional value θc.

FIG. 2 specifically illustrates the vector control calculating part 9 a applied to the torque controller in FIG. 1.

The d-axis current control part 9 a 1 in FIG. 2 outputs a second commanded current Id** so that the d-axis detected current Idc matches (follows) the d-axis commanded current Id*. Similarly, the q-axis current control part 9 a 2 outputs a second commanded current Iq** so that the q-axis detected current Iqc matches (follows) the q-axis commanded current Iq*. The d-axis current control part 9 a 1 and q-axis current control part 9 a 2 perform proportional integration or only integration and determine their control gain from a control response angular frequency to be set in current control.

A vector calculating part 9 a 3 calculates the commanded voltages Vdc* and Vqc* for the d-axis and q-axis by using Id** output from the d-axis current control part 9 a 1 and Iq** output from the q-axis current control part 9 a 2, the calculated frequency col, and motor constants as in equation (1) so as to control the output voltage V₁ from the electric power converter 2.

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack & \; \\ \left. \begin{matrix} {V_{dc}^{*} = {{{Id}^{**} \cdot R^{*}} - {\omega_{1} \cdot {Lq}^{*} \cdot \frac{1}{1 + {{Tq} \cdot s}} \cdot {Iq}^{**}}}} \\ {V_{qc}^{*} = {{{Iq}^{**} \cdot R^{*}} + {\omega_{1} \cdot {Ld}^{*} \cdot \frac{1}{1 + {{Td} \cdot s}} \cdot {Id}^{**}} + {\omega_{1} \cdot {Ke}^{*}}}} \end{matrix} \right) & (1) \end{matrix}$

Where R* is a resistance setting, Ld* is a d-axis inductance setting, Lq* is a q-axis inductance setting, Ke* is an induced voltage constant setting, Tq equals Lq*/R*, Td equals Ld*/R*, and s is a Laplace operator.

The output voltage calculating part 10 calculates the output voltage V₁ by using the commanded voltages Vdc* and Vqc*, which are output from the vector control calculating part 9 a, as in equation (2).

[Equation 2]

V ₁=√{square root over (V _(dc)*² +V _(qc)*²)}  (2)

First, the commanded torque correcting part 7, which is a feature of the present invention, will be described.

FIG. 3 shows the structure of the commanded torque correcting part 7 in the first embodiment of the present invention. The commanded torque correcting part 7 outputs a compensation amount Δτ* to lower a commanded torque τ* when the output voltage V₁ of the electric power converter 2 reaches the prescribed voltage V₁ ^(*) _(ref).

An integration gain calculating part 71 calculates an integration gain Ka by using the calculated frequency ω₁ and constants (R and Lq) as in equation (3).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack & \; \\ {{Ka} = \frac{\omega \; c}{\sqrt{R^{*2} + \left( {\omega_{1} \cdot L_{q}^{*}} \right)^{2}}}} & (3) \end{matrix}$

Where ωc is the control response angular frequency, R* is a resistance setting of the motor and harness, and Lq* is the q-axis inductance of the motor.

In equation (3), if the motor or the frequency range satisfies R*²<<(ω₁L_(q)*)², R*² can be omitted in the calculation of equation (3). The commanded torque correcting part 7 is an outer loop of the vector control calculating part 9 a, which controls current; the control response angular frequency ωc is preferably about half to one-tenth the control response angular frequency to be set in the current control circuit to obtain stable control.

A multiplier 72 calculates the integration gain Ka, which is output from the integration gain calculating part 71, and the difference between the prescribed voltage V₁*_(ref) and output voltage V₁ from the electric power converter 2. The calculation result is output to a limited integration calculating part 73.

The limited integration calculating part 73 performs an integration calculation. In this calculation, positive values are limited to 0, and negative values are limited to a value obtained by multiplying the absolute value of an uncorrected commanded torque τo*, which is given by the high-end device, by −1. That is, correction to the commanded torque τ*, which has a polarity opposite to that of the commanded torque τo* given by the high-end device, does not occur, and thereby an inverted operation is not unintentionally performed.

τo* is also input to a polarity determining part 74. The polarity determining part 74 outputs a polarity signal Sign [τo*], which is +1 or −1, according to equation (4).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 4} \right\rbrack & \; \\ \left. \begin{matrix} {{{\tau_{0}^{*} \geq 0}:{{Sign}\left\lbrack \tau_{0}^{*} \right\rbrack}} = {+ 1}} \\ {{{\tau_{0}^{*} < 0}:{{Sign}\left\lbrack \tau_{0}^{*} \right\rbrack}} = {- 1}} \end{matrix} \right) & (4) \end{matrix}$

A multiplier 75 receives an integrated value output from the limited integration calculating part 73 and the polarity signal, which is +1 or −1, output from the polarity determining part 74. These values are used to correct the torque command with the inverted polarity of the commanded torque τo* given by the high-end device. Specifically, the polarity of Δτ* is determined according to equation (5).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 5} \right\rbrack & \; \\ \left. \begin{matrix} {{\tau_{0}^{*} \geq 0}:{{\Delta \; \tau^{*}} \leq 0}} \\ {{\tau_{0}^{*} \leq 0}:{{\Delta\tau}^{*} \geq 0}} \end{matrix} \right) & (5) \end{matrix}$

Finally, the corrected commanded torque Δτ is calculated as in equation (6).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack & \; \\ {{\Delta\tau}^{*} = {\frac{K}{s} \cdot \left( {V_{ref}^{*} - V_{1}} \right) \cdot {S_{ign}\left\lbrack \tau_{0}^{*} \right\rbrack}}} & (6) \end{matrix}$

In addition, the corrected commanded torque Δτ* is added to the commanded torque τo*, calculating a new commanded torque τ*, as in equation (7).

[Equation 7]

τ*=τ₀*+Δτ*  (7)

Referring to FIG. 1 again, the current command converting part 8 calculates the commanded currents Id* and Iq* for the d-axis and q-axis by using the commanded torque τ*, which has been corrected, as in equation (8), and outputs the calculated values.

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack & \; \\ \left. \begin{matrix} {{Id}^{*} = 0} \\ {{Iq}^{*} = \frac{\tau^{*}}{K_{t}}} \end{matrix} \right) & (8) \end{matrix}$

Although Id* is set to 0, if an operation is performed with a weak magnetic field or the maximum torque-to-current ratio, Id* may be a prescribed value rather than 0.

FIG. 4 illustrates control characteristics under no load when a calculation for torque command correction is performed according to the first embodiment. The commanded torque correcting part 7 needs to set the prescribed voltage V₁*_(ref) at which a calculation for torque command correction starts.

When the electric power converter 2 undergoes sine wave modulation, the average of its maximum output voltage is represented by equation (9), so V₁*_(ref) must be set to a value slightly lower than the value calculated according to equation (9).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 9} \right\rbrack & \; \\ {\overset{\_}{V_{1\; \max}^{*}} = {\frac{3}{4} \cdot {Ed}}} & (9) \end{matrix}$

The setting must satisfy the relation in equation (10).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 10} \right\rbrack & \; \\ {{V_{1}^{*}{ref}} < {\frac{3}{4} \cdot {Ed}}} & (10) \end{matrix}$

Where Ed is a DC voltage.

When an in-phase signal (a three-fold harmonic component, for example), which is canceled at a line voltage, is superimposed to the commanded three-phase voltages Vu*, Vv*, and Vw*, the average of the maximum output voltage of the electric power converter 2 can be increased. The average of the maximum output voltage is represented by equation (11).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack & \; \\ {\overset{\_}{V_{1\; \max}^{*}} = {\frac{\sqrt{3}}{2} \cdot {Ed}}} & (11) \end{matrix}$

The setting must satisfy the relation in equation (12).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 12} \right\rbrack & \; \\ {{V_{1}^{*}{ref}} < {\frac{\sqrt{3}}{2} \cdot {Ed}}} & (12) \end{matrix}$

Effects of the first embodiment will be described next with reference to FIGS. 4 and 5.

When the commanded torque τo* is given from the high-end device at point A in time, the rotational speed ωr of the permanent magnet synchronous motor 1 increases and the motor generates an induced voltage, increasing the output voltage V₁ from the electric power converter 2.

The output voltage V₁ reaches the prescribed voltage V₁*_(ref) at point B in time, so correction of τ* starts at that time.

Since there is no load at time C in time, τ* is corrected to approximately 0 and ωr is fixed at a constant value, indicating that a stably controlled operation is achieved.

Operation continues in this state. When a load torque τL is applied at point D in time, ωr drops. However, a torque τm is generated by an amount equal to τL, indicating that a stably controlled operation is achieved at a point above the equivalent point of the N-T (rotational speed-torque) characteristics.

In the first embodiment, the integration gain Ka is calculated in the integration gain calculating part 71 according to equation (3) so that the response time of the commanded torque correcting part 7 becomes constant regardless of the rotational speed or of the motor. However, in a case (or an application) in which the output voltage V₁ may cause a slight overshoot or the response time may be delayed, the integration gain Ka may be a constant value.

Since, in the first embodiment, the output voltage V₁ is used to correct the commanded torque τ*, if the resistance R changes, V₁ is affected by the change; the larger the resistance R is, the larger V₁ is. This indicates that since V₁, which is affected by a change in the resistance R, is controlled, the torque controller is robust against the resistance R.

Even if the resistance R, which is used in integration gain calculation according to equation (3), slightly changes, quick response is slightly lowered but stability is not largely affected.

Although the vector control calculating part 9 a has been used in the first embodiment, the vector control calculation part 9 b shown in FIG. 6 may be used instead.

The d-axis current control part 9 b 1 outputs a compensated voltage ΔVd so that the d-axis detected current Idc matches (follows) the d-axis commanded current Id*. Similarly, the q-axis current control part 9 b 2 outputs a compensated voltage ΔVq so that the q-axis detected current Iqc matches (follows) the q-axis commanded current Iq*. The d-axis current control part 9 b 1 and q-axis current control part 9 b 2 perform proportional integration or only integration and determine their control gain from a control response angular frequency to be set in current control. A vector calculating part 9 b 3 calculates the commanded voltages Vdc* and Vqc* for the d-axis and q-axis by using AVd and AVq output from the current control parts 9 b 1 and 9 b 2, the calculated frequency ω₁, and the motor constants as in equation (13) so as to control the output voltage V₁ from the electric power converter 2.

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack & \; \\ \left. \begin{matrix} {V_{dc}^{*} = {{{Id}^{*} \cdot R^{*}} - {\omega_{1} \cdot {Lq}^{*} \cdot \frac{1}{1 + {{Tq} \cdot s}} \cdot {Iq}^{*}} + {\Delta \; V_{d}}}} \\ {V_{qc}^{*} = {{{Iq}^{*} \cdot R^{*}} + {\omega_{1} \cdot {Ld}^{*} \cdot \frac{1}{1 + {{Td} \cdot s}} \cdot {Id}^{*}} + {\omega_{1} \cdot {Ke}^{*}} + {\Delta \; {Vq}}}} \end{matrix} \right) & (13) \end{matrix}$

A vector control calculation part 9 c shown in FIG. 7 may be used instead of the vector control calculation parts 9 a and 9 b.

The d-axis current control part 9 c 1 outputs the d-axis commanded voltage Vdc* so that the d-axis detected current Idc matches (follows) the d-axis commanded current Id*. Similarly, the q-axis current control part 9 c 2 outputs the q-axis commanded voltage Vqc* so that the q-axis detected current Iqc matches (follows) the q-axis commanded current Iq*. The d-axis current control part 9 c 1 and q-axis current control part 9 c 2 perform proportional integration or only integration and determine their control gain from a control response angular frequency to be set in current control. The d-axis commanded voltage Vdc* and the q-axis commanded voltage Vqc* are used to control the output voltage V₁ from the electric power converter 2.

Calculations in the vector control calculation parts 9 b and 9 c, as described above, can provide the same effect as in the vector control calculation part 9 a.

Second Embodiment

FIG. 8 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor in a second embodiment of the present invention.

This embodiment is applied to a torque controller from which a position sensor such a resolver or an encoder is eliminated. In FIG. 8, the same components as in FIG. 1 are denoted by the same reference characters. An axial error calculating part 12 infers an axial error Δθc (=θc′−θ), which is a difference between an inferred phase value θc′ and motor phase value θ by using the d-axis commanded voltage Vdc*, the q-axis commanded voltage Vqc*, the detected currents Idc and Iqc, an inferred frequency ω₁′, and the motor constants, as in equation (14).

$\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack & \; \\ {{{\Delta\theta}\; c} = {\tan^{- 1}\left\lbrack \frac{{Vdc}^{*} - {R^{*} \cdot {Idc}} + {\omega_{1}^{\prime} \cdot {Lq}^{*} \cdot {Iqc}}}{{Vqc}^{*} - {R^{*} \cdot {Iqc}} - {\omega_{1}^{\prime} \cdot {Lq}^{*} \cdot {Idc}}} \right\rbrack}} & (14) \end{matrix}$

A frequency inferring part 13 calculates the inferred frequency ω₁′ so that the calculated axial error Δθc becomes 0. A phase calculating part 14 integrates the inferred frequency ω_(c)′ to calculate the inferred phase value θc′.

Even in an inexpensive control system of this type, in which a position sensor is eliminated, it is clear that the same operation as in the first embodiment is achieved and the same effect can be obtained.

Although the vector control calculating part 9 a is used in the second embodiment, the use of the vector control calculating part 9 b or 9 c described in the first embodiment can provide the same effect as in the second embodiment.

Third Embodiment

Although the expensive current detector 3 is used in the first and second embodiments to detect the three-phase AC currents iu, iv, and iw, the present invention can also be applied to a torque controller that detects currents in an inexpensive manner.

FIG. 9 shows the structure of a torque controller in a permanent magnet synchronous motor according to a third embodiment of the present invention, the torque controller detecting currents as described above. In FIG. 9, the same components as in FIG. 8 are denoted by the same reference characters. A current inferring part 15 infers the three-phase currents iu, iv, and iw, which flow in the permanent magnet synchronous motor 1, from a DC current I_(DC) flowing in the input bus of the electric power converter 2.

The coordinate converting part 6 uses inferred currents iû, iv̂, and iŵ to calculate the detected currents Idc and Iqc for the d-axis and q-axis.

Even in an inexpensive control system of this type, in which a current sensor is eliminated, it is clear that the same operation as in the first embodiment can be achieved and the same effect can be obtained.

Although the vector control calculating part 9 a is used in the third embodiment, the use of the vector control calculating part 9 b or 9 c described in the first embodiment provides the same effect as in the third embodiment.

FIG. 10 is a block diagram showing the structure of a torque controller for a permanent magnet synchronous motor when the present invention is applied to an electric hydraulic pump system.

In this example, the third embodiment of the present invention is applied to an electric hydraulic pump system mounted on a vehicle.

In FIG. 10, the same components as in FIG. 9 are denoted by the same reference characters.

A hydraulic pump system 22, which includes the permanent magnet synchronous motor 1, is driven by an inverter controller 23.

The components denoted by reference characters 2, 6 to 8, 9 a, 10 to 15, and 21 in FIG. 10 are implemented by software and hardware circuits.

When the third embodiment of the present invention is applied to a hydraulic pump system as described above, quick-response and high-precision control characteristics can be achieved.

Although the third embodiment is used in this example, the first or second embodiment may be used instead. Furthermore, even when the vector control calculating part 9 b or 9 c described in the first embodiment may be used instead of the vector control calculating part 9 a, the same effect as in this example can be obtained.

With the torque controller for a permanent magnet synchronous motor that embodies the present invention, when the output voltage of the electric power converter 2 reaches the prescribed voltage V₁*_(ref), a motor torque or motor current is generated according to the load torque.

From the control characteristics according to the first embodiment, shown in FIGS. 4 and 5, it is found that when there is no load torque, a new commanded torque τ* converges to 0 and that when there is a load torque, the new commanded torque τ* becomes equivalent to the load torque.

According to the embodiments described above, a vector controller for a permanent magnet synchronous motor can achieve highly precise torque control at a quick response even near a point at which the output voltage of an electric power converter is limited (saturated).

In addition, a torque controller for a permanent magnet synchronous motor that can be applied to both an inexpensive current detecting system and a system from which a position sensor is eliminated can be provided. 

1. A torque controller for a permanent magnet synchronous motor comprising: an electric power converter for driving the permanent magnet synchronous motor; a current command calculator for calculating two current command values for a q-axis (torque axis) and a d-axis (magnetic flux axis) based on a torque command value; and a vector controller for creating two output voltage command values of the electric power converter for the q-axis and the d-axis based on two differences between the current command value and a detected current value for the q-axis and the d-axis, respectively, to control an output voltage value of the electric power converter; wherein the torque command value is corrected so that the output voltage of the electric power converter does not exceed a prescribed voltage value.
 2. A torque controller for a permanent magnet synchronous motor comprising: an electric power converter for driving the permanent magnet synchronous motor; a current command calculator for calculating two current command values for a q-axis (torque axis) and a d-axis (magnetic flux axis) based on a torque command value; and a vector controller for creating two output voltage command values of the electric power converter for the q-axis and the d-axis based on two differences between the current command value and a detected current value for the q-axis and the d-axis, respectively, to control an output voltage value of the electric power converter; wherein when the output voltage value of the electric power converter reaches a prescribed voltage value, an integration is performed on a difference between the prescribed voltage value and the output voltage value of the electric power converter to obtain an integrated value and the integrated value is used to correct a commanded torque value.
 3. The torque controller according to claim 1, wherein the prescribed voltage value is lower than a saturation voltage of the electric power converter.
 4. The torque controller according to claim 1, wherein, in correction of the commanded torque value, an integration is performed on a difference between the prescribed voltage value and the output voltage value of the electric power converter to obtain an integrated value, the integrated value is multiplied by a sign of a polarity of the commanded torque value before the correction to obtain a multiplied value, and the multiplied value is added to the commanded torque value before the correction.
 5. The torque controller according to claim 1, wherein, in correction of the commanded torque value, which is denoted τ*, calculations as in the equations shown below are performed by using the prescribed voltage value, which is denoted V_(ref)*, and the output voltage value of the electric power converter, which is denoted V₁. $\begin{matrix} \left\lbrack {{Equation}\mspace{14mu} 1} \right\rbrack & \; \\ {{Ka} = \frac{\omega \; c}{\sqrt{R^{*2} + \left( {\omega_{1} \cdot L_{q}^{*}} \right)^{2}}}} & (3) \\ \left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack & \; \\ {{\Delta\tau}^{*} = {\frac{K}{s} \cdot \left( {V_{ref}^{*} - V_{1}} \right) \cdot {S_{ign}\left\lbrack \tau_{0}^{*} \right\rbrack}}} & (6) \\ \left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack & \; \\ {\tau^{*} = {\tau_{0}^{*} + {\Delta \; \tau^{*}}}} & (7) \end{matrix}$ where τo* is the command torque value before the correction, Δt is an amount of torque correction, τ* is a commanded torque value after the correction, Ka is an integrated gain, ω₁ is a calculated frequency value, ωc is a control response angular frequency, R is the resistance of a motor and a harness, Lq is an inductance value, and Sign[τo*] is ±1.
 6. The torque controller according to claim 2, wherein a gain in the integration is a control response angular frequency to be set in a commanded torque correcting part, the control response angular frequency being about half to one-tenth the control response angular frequency to be set in current control by a vector control calculating part.
 7. The torque controller according to claim 2, wherein an upper limit of the integrated value calculated from the difference between the prescribed voltage value and the output voltage value of the electric power converter is restricted to 0, and a lower limit of the integrated value is restricted to a value obtained by multiplying the absolute value of the commanded torque value before the correction by −1.
 8. The torque controller according to claim 1, wherein the prescribed voltage value is three-fourths or less of a DC voltage supplied to the electric power converter when sine wave modulation is performed on commanded voltage values Vu*, Vv*, and Vw* of a three-phase AC current, which control the electric power converter, or 2/√{square root over (3)} or less of the DC voltage supplied to the electric power converter when an in-phase signal (a three-fold harmonic component, for example), which is canceled at a line voltage, is superimposed on the commanded voltage values Vu*, Vv*, and Vw* of the three-phase AC current.
 9. A torque controller for a permanent magnet synchronous motor, the torque controller serving as a vector controller that performs torque control to control an output voltage value of an electric power converter that drives the permanent magnet synchronous motor, wherein the torque controller generates a motor torque or a motor current according to a load torque when the output voltage value of the electric power converter reaches a prescribed voltage value.
 10. The torque controller according to claim 2, wherein the calculated frequency value is obtained from commanded voltage values for the d-axis and q-axis, and a detected motor current or a reproduced motor current so that a difference calculated from an inferred phase value and a motor phase value becomes
 0. 11. The torque controller according to claim 1, wherein the detected current values are used to reproduce a motor current according to a detected input DC bus current value of the electric power converter.
 12. A torque controller for a permanent magnet synchronous motor, wherein the controller described in claim 1 is applied to an electric hydraulic pump system. 